Current reduction by dynamic receiver adjustment in a communication device

ABSTRACT

A method for reducing current drain in a communication device includes detecting ( 602, 604 ) interferers outside of a receiver passband of the communication device, measuring power levels and frequency offsets ( 608, 808 ) of the interferers and a transmitter of the communication device, and determining ( 609, 809 ) whether crossmodulation products exceed a noise spectrum threshold within the receiver passband. If the crossmodulation products exceed the threshold, the method includes calculating ( 605, 805 ) a receiver linearity required to achieve a desired signal-to-interference ratio and adjusting the receiver linearity calculated in the calculating step to achieve the desired signal-to-interference ratio.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a divisional of pending U.S. application Ser. No.10/630,124 filed Jul. 30, 2003, and assigned to Motorola, Inc., fromwhich priority is hereby claimed under 35 U.S.C. § 120.

FIELD OF THE INVENTION

The present invention generally relates to reducing power consumption ina communication device such as a radiotelephone. More particularly, thepresent invention relates to a technique for dynamic adjustment ofreceiver elements in radiotelephone device.

BACKGROUND OF THE INVENTION

Code Division Multiple Access (CDMA) and Wideband CDMA (WCDMA) receiversin North American bands have been and will continue to coexist withother narrow band systems such as AMPS, IS136, and Global System forMobile (GSM) communication system. This situation leads to narrow bandinterference signals that cause both crossmodulation and intermodulationnon-linear distortions. The traditional method to mitigate thedistortion is to operate with sufficiently high linearity at the expenseof additional current drain. Additionally, receiver designs aremigrating to higher dynamic range analog-to-digital converters (A/Ds)with digital filters and less analog filtering. This type of receiverdesign therefore passes both the desired signal and interference througha set of high dynamic range circuit elements before final elimination ofthe interference by digital filters. Again, the dynamic range of thesecircuit elements (A/Ds and filters) is set sufficiently high toaccommodate the largest expected interference at the expense of currentdrain and battery life. These two factors cause receivers to operatewith higher current drain than needed under the majority of usageconditions.

Interference is a particular problem in CDMA systems which require thereceiver and transmitter to be on continuously. An example of such aspecification is Telecommunications Industry Association/ElectronicIndustry Association (TIA/EIA) Interim Standard IS-95, “MobileStation-Base Station Compatibility Standard for Dual-Mode WidebandSpread Spectrum Cellular System” (IS-95). IS-95 defines a directsequence code division multiple access (DS-CDMA or CDMA) radiotelephonesystem. In a CDMA system, the receiver must be on continuously in orderto receive incoming data while on a traffic channel and the transmittermust be on continuously while on a traffic channel.

Prior art receiver linearity systems dynamically adjust the linearity ofa receiver based on detection of poor received signal quality based oncarrier-to-interference ratio (Ec/Io) or FER (frame error rate),knowledge of the transmit level, and received signal strengthindications (RSSI). Also, the prior art captures the various ways thatlinearity can be changed, i.e. either gain change or current change inreceiver amplification stages. However, the prior art does not make themost efficient use of current drain since the linearity is potentiallyincreased without any knowledge of the interference. Poor quality can becaused by a number of factors that are not related to receiver linearityand thus is not the most efficient metric. Furthermore, the prior artdoes not address the use of variable dynamic range in the baseband.

In another technique, the quality metric shortcoming is somewhataddressed by performing a spectral estimation of the in-band signal.This spectral estimation is performed in post-channel filtering to lookfor narrowband intermodulation distortion products and a low-noiseamplifier (LNA) bypass is used to improve receiver linearity. Note thatthis addresses potential intermodulation products but does not considercrossmodulation. Also, the mitigation is limited to LNA bypass and thuslimits the optimization of performance near sensitivity. Furthermore,detection of the intermodulation products is much more difficult thandetection of the interference since the intermodulation products areseveral dB lower in amplitude than the actual interference.

Dynamic control of A/D converter dynamic range is also known, wherein ananalog detected voltage is used to vary an A/D's dynamic range. Thevariable control of converter range is often internal to the A/D and isnot part of a larger system. This limits the efficacy of the techniquesto mitigation of narrow-band interferer by the A/D and does not providea mechanism to reduce the current drain of the other circuit elements,e.g. digital filters, analog filters, and RF circuits.

Accordingly, there is a need for a method and apparatus for reducingcurrent drain in a communication device such as a radiotelephone whenmitigating non-linear distortion effects. There is a further need toreduce the current drain by the receiver in a communication deviceoperating in a CDMA system, without sacrificing the ability to receiveincoming signals. It would also be of benefit to provide theseadvantages without additional hardware, which would increase the cost ofthe communication device.

BRIEF DESCRIPTION OF THE DRAWINGS

The features of the present invention, which are believed to be novel,are set forth with particularity in the appended claims. The invention,together with further objects and advantages thereof, may best beunderstood by making reference to the following description, taken inconjunction with the accompanying drawings, in the several figures ofwhich like reference numerals identify identical elements, and wherein:

FIG. 1 shows a block diagram of a prior art multimode communicationdevice;

FIG. 2 is a flow chart illustrating the operation of the device of FIG.1;

FIG. 3 shows a block diagram of a first embodiment of a multimodecommunication device, in accordance with the present invention;

FIG. 4 is a flow chart illustrating the operation of the device of FIG.3;

FIG. 5 shows a block diagram of an alternate embodiment of a multimodecommunication device, in accordance with the present invention;

FIG. 6 is a flow chart illustrating the operation of the device of FIG.5; and.

FIG. 7 is a flow chart for a method of reducing current drain in areceiver, in accordance with the present invention.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT

The present invention provides a unique method to improve reception in areceiver circuit for a wireless communications device in a communicationsystem while reducing current drain. This is done by reducing thecurrent drain in the receiver by making the elements in the receiver(mixers, amplifiers and filters) less linear when there is little or nointerference, and increasing linearity again when interference exists.In particular, the present invention reduces the effects ofintermodulation distortion and crossmodulation distortion by directdetermination of the interference, instead of relying on channelquality, and by improving linearity of the receiver only when non-lineardistortion products are anticipated. This improvement is accomplishedwithout any significant additional hardware or cost in the communicationdevice. Moreover, the present invention can take advantage of existinghigh dynamic range devices such as analog-to-digital converters anddigital filters in combination with solutions for processing of the RFinput signals necessary in the communication device.

Specifically, the present invention resolves the problems of the priorart by using in-band and out-of-band interference detection and spectrumestimation to identify the power level and frequency offsets ofinterference to decide if linearity reduction is performed, rather thanusing a CDMA channel quality metric or for use in a GSM environment(e.g. for cross modulation or intermodulation issues). Further, thepresent invention can also utilize dynamic range reduction to the sameeffect and linearity adjustments, wherein dynamic range adjustments andlinearity adjustments can be used separately or in combination.

Turning to FIG. 1, a block diagram of a prior art communication deviceis shown. Typically, this device is a duplex CDMA cellularradiotelephone operable in a CDMA radiotelephone system operatingaccording to TIA/EIA Interim Standard IS-95, “Mobile Station-BaseStation Compatibility Standard for Dual-Mode Wideband Spread SpectrumCellular System,” operating at 800 MHz. Alternatively, theradiotelephone system 100 could operate in accordance with other CDMAsystems including PCS systems at 1800 MHz or with any other suitabledigital radiotelephone systems. The device includes an antenna 106coupled to a transmitter 100 and a receiver circuit 102, which cancommunicate simultaneously, although in different frequency bands,through a duplexer filter 104 with an antenna 106. The device iscontrolled by one or more of a microprocessor (not shown),microcontroller, or digital signal processor 108 (DSP), which generatethe necessary communication protocol for operating in a compatiblecellular system and can perform many other devices that providefunctions for the wireless communication device, such as writing to adisplay, accepting information from a keypad, communicating through auser interface, audio, etc. These other devices are not shown forsimplicity of the figures and to avoid confusion. A battery 105 providesoperating power to the other components of the radiotelephone.Preferably, the battery is rechargeable.

The antenna 106 receives RF signals from a base station 102 in thevicinity. The received RF signals are converted to electrical signals bythe antenna 106 and provided to the receiver 102, which providesconversion to baseband signals. The receiver 102 includes an amplifierand other circuitry, such as RF circuits and demodulation circuitry, asis known in the art. The baseband signals are provided to the othercircuits (not shown) in the radiotelephone, which converts them tostreams of digital data for further processing.

Similarly, the radiotelephone provides baseband signals throughmodulation circuitry to the transmitter 100, which sends electrical RFsignals to the antenna 106 for transmission to the base station andother base stations in the vicinity. The amplifier 126 typicallyconsumes the most current drain while on a traffic channel althoughtechniques exist to reduce this at lower transmit levels. At lowertransmit levels, the current consumption of the various receiver 102elements becomes more critical to the radio current drain. Additionally,the current drain while monitoring a paging channel is greatly impactedby the current drain of receiver 102 since the amplifier 126 andtransmitter 100 are turned off.

The control circuitry, such as DSP 108, controls functions of theradiotelephone 104. The control circuitry operates in response to storedprograms of instructions and includes a memory (not shown) for storingthese instructions and other data. The control circuitry is also coupledto other elements of the radiotelephone, which are not shown so as tonot unduly complicate the drawing. For example, the radiotelephone willtypically include a user interface to permit user control of theoperation of the radiotelephone. The user interface typically includes adisplay, a keypad, a microphone and an earpiece. The user interface iscoupled to the control circuitry.

The DSP 108 can control receiver linearity by adjusting a gain stage 110and mixer 118 through a gain control 112 or a current control 114. Thereceiver circuit path also includes interstage filtering 116, analogfiltering 120, and analog-to-digital conversion 122 for the receiverbackend 124, as are generally known in the art. The DSP 108 determines areceived signal strength indication (RSSI) for the receiver channel. Itshould be recognized that RSSI can be determined in various separateblocks or combinations thereof (such as through an AGC system). Thereceiver backend 124 includes a demodulator, signal processing, andother circuitry known in the art to perform baseband conversion andappropriate active filtering which is necessary for demodulation of thedesired communication signal. Also, the receiver backend 124 can usedigital processing to determine a quality of the receiver channel (i.e.frame error rate (FER), Ec/Io (carrier-to-interference ratio), and thelike).

The gain stage 110 is a pre-amp that uses automatic gain control (AGC)to control the signal gain input to the backend baseband circuit(demodulator) 124, since this circuit is susceptible to overload. TheAGC maintains each stage's power level within the designed operatingrange so the receivers may function properly. The receiver circuit canbe direct conversion or have one or more intermediate frequency stages.The mixer 118 converts the signal into a baseband representation whichis then subsequently filtered 120 by a baseband filter that allowsprimarily the desired communication signal to pass for furtherprocessing. Although filtered, the signal, plus noise, interference andintermodulation are present. Some of these signals are on the desiredcommunication signal frequency band, such as intermodulation productsgenerated due to crossmodulation with the co-transmitting poweramplifier 126, and are passed on to further processing. After the filter120, the signal is converted to a digital signal by theanalog-to-digital converter 122. This converter takes all signals(desired communication signal and interference) and converts them todigital data bits which are then further processed including additionalfiltering and demodulation.

The digital signal processor 108 comprises a detector for detectingon-band interference from all sources including self-interference fromthe transmitter power amplifier 126 as well as external channel noise.The detector estimates the channel quality and the received signalstrength to determine the proper amount of gain or current to supply tothe gain stage 110 and mixer 118 to adjust the linearity thereof forreduced crossmodulation. It should be recognized that many of thesestages can be incorporated solely in software without need for hardwareembodiments.

FIG. 2 shows a prior art technique for linearity adjustment forintermodulation reduction, used with the device of FIG. 1. The devicefirst determines a received channel quality 200 using techniques knownin the art. If the channel quality is good 202 then the effect ofintermodulation and other noise sources are not significant and thedevice can operate nominally 204. However, if the channel quality ispoor then intermodulation can be one of the causes. In this case, thetransmitter power amplifier level is used and compared against a firstthreshold 208 that is empirically determined and is a function of systemdesign. If the power level does not exceed the first threshold then theinterference or distortion will not be anticipated as significant andthe device can operate nominally 204. However, if the transmitter powerlevel exceeds the threshold then crossmodulation can be the cause, andthe prior art then calls for a linearity adjustment of the low-noiseamplifier (LNA) and mixer.

What type of linearity adjustment to be used is decided by determining210 a received signal strength indication (RSSI) and compared the RSSIwith a second empirically determined threshold 212 to determine the typeof operating condition adjustment (i.e. gain or linearity). Inparticular, if the RSSI is greater than the second threshold then thereis sufficient signal present and a gain reduction would not furtherreduce the existing poor signal in relation to the interference, i.e.signal-to-interference ratio (S/I). In other words, if the signal weresmall (small RSSI), then S/(N+I) would be the proper representation, asa result of a gain reduction. The signal and interference would now besmaller, while noise is a constant and relatively larger. However, thedistortion products within the receiver are reduced greatly by loweringa gain 216 of the gain stage 110 and mixer 118 of the receiver thusimproving the existing poor signal in relation to the interference.Lowering the gain is preferred as current dissipation is typicallyreduced for reduced gain. However, if the RSSI is below or equal to thesecond threshold 212 (i.e. not enough signal) then lowering the gainwould incur a more negative penalty, since there is not enough signal inthe first place, and the linearity increasing adjustment is accomplishedby increasing current 214 to the gain stage of the receiver.

The present invention is different in several respects from the aboveprior art. First, channel quality is not used for an intermodulation ora crossmodulation determination since channel quality can be degraded bymany other causes than intermodulation or crossmodulation. Second, afrequency offset of the interference from the receiver band isdetermined and used in the linearity determination. And third, the powerlevels of the interference are used in evaluating whether to adjustlinearity.

The present invention defines a method to reduce the receiver currentdrain during periods in which there is little or no interference, thusreducing the overall current consumption for the communication device.To operate within the IS-95 standard, a CDMA receiver must meet thespecifications for two-tone intermodulation (IM) and single-tonedesensitization (STD) defined in TIA/EIA-98. These two requirements setthe linearity requirement for the receiver front end, which in turn setsits power consumption. The IM specification requires that the receiverfront end be linear enough to reduce the level of on-channel third-orderintermodulation product of two equally spaced continuous waveinterferers. Further, the STD specification requires that the receiverfront end be linear enough to reduce the level of an on-channelcross-modulation product of a continuous wave interferer and the radio'sown transmit signal.

In the present invention, the linearity of the receiver 102 iscontrolled by a linear adjust signal from the control circuitry 108. Thelinear adjust signal controls the current flow to the receiver 102, andin particular the bias current to the receiver amplifier and mixer. Thecontrol circuitry 108 can also provide a gain adjust signal to thereceiver 102. The gain adjust is independent of the linearity adjust anddoes not depend on current limiting. The gain required in the receiveris independent of whether the communication device is transmitting. Athigh gain, the receiver is more susceptible to interference and controlof the receiver linearity becomes important, as described previously.Increasing the receiver linearity requires the receiver to use morecurrent.

Turning to FIG. 3, a block diagram is shown of a communication device,such as are operable in GSM and WCDMA communication systems, inaccordance with the present invention. Preferably, this device is acellular radiotelephone incorporating the present invention. The deviceincludes a transceiver including a transmitter 100 as before and areceiver circuit 302, which can communicate simultaneously, although ondifferent frequency bands, through a duplexer filter 104 with an antenna106. The device is controlled by one or more of a microprocessor (notshown), microcontroller, or digital signal processor (DSP) module 308(which can be a portion of the device DSP), which generates thenecessary communication protocol for operating in a compatible cellularsystem and for performing many other functions for the wirelesscommunication device, such as writing to a display, acceptinginformation from a keypad, communicating through a user interface,audio, etc. (not shown for the simplicity). The DSP module 308 cancontrol receiver linearity by adjusting its gain stage 310, mixer 318and analog filter block 320 through a gain control 112 or a currentcontrol 114. The receiver includes a mixer 318, analog filtering block320, an analog-to-digital converter (ADC) 322 for the receiver backend324 and a digital filter block 323 after the ADC 322.

The present invention makes novel use of the high dynamic range, widebandwidth ADC 322 that is different from the prior art since it allowsout-of-band and high-level signal detection in addition to in-bandsignal detection. The DSP module 308 can determine out-of-band signalsincluding interference, which are passed by the ADC 322. The powerlevels and frequency offsets of the interferers are ultimately used todetermine linearity adjustments, as will be described below. The DSPmodule is in effect a detector for estimating non-linear distortionproducts due to intermodulation of multiple interferers orcrossmodulation of the leakage from the device's PA 126 with aninterferer. The detector can use the in-band and out-of-band signalproducts, as well as transmitter power level information from the PA 126of the duplex transceiver, to determine the proper linearity adjustmentfor the gain stage(s) 310 and mixer(s) 318 and proper dynamic range forthe A/D 322 and filter 323 for reduced distortion. Based on theinterferer measurements and optionally the transmitter power level asignal-to-interference ratio can be calculated, wherein if thesignal-to-interference ratio does not exceed a predetermined threshold,then the linearity of the receiver is increased using either increasedcurrent or gain change and/or dynamic range, as will be explained below.

FIG. 4 shows a graph illustrating the effects of third-orderintermodulation on a receiver (RX) band. In this example, a firstinterferer, A₁ cos(ω₁t), which is nearer to the RX band intermodulateswith a second interferer, A₂ cos(ω₂t), which is farther from the RXband. This intermodulation produces third-order products within the RXband that degrades signal-to-interference performance (as well asthird-order products outside of the band). In particular, thethird-order interference isν_(imd3)(t)=¾A ₁ ² A ₂ cos[(2ω₁−ω₂)t]+¾A ₂ ² A ₁ cos[(2ω₂−ω₁)t]  (1)and the signal-to-interference ratio is $\begin{matrix}{\frac{P_{sig}}{P_{imd}} = \frac{P_{sig}}{{2P_{1}} + P_{2} - {2{IP}_{3}}}} & (2)\end{matrix}$where P_(sig) is the power of the desired in-band (RX) signal andP_(imd) is the power of the interference. P₁ is the power level of thenearer interferer, and P₂ is the power level of the farther interferer.In the present invention, the power and frequency offsets of theinterferers are measured directly to determine their effect within theRX band. With a given signal-to-interference ratio (SIR) limit (asdefined in the standard), an IP₃ threshold can then be found to providesufficient SIR. The components of the receiver can then have theirlinearity adjusted to provide sufficient SIR.

FIG. 5 shows the DSP module (308 in FIG. 3) as used to detectinterference and estimate intermodulation distortion, in accordance withthe present invention. In this case, the non-linear distortion that isproduced is in a relatively narrowband. The DSP uses inputs of themeasured transmitter power (TX) and the received signals level (P_(sig))as well as the required SIR. Then, given the measurements of theinterferers, can calculate an IP₃ threshold from equation 2, to producethe desired SIR. In particular, the present invention uses spectralestimation to decide if any significant intermodulation products fromnarrowband interference will fall inside the receive channel filter. Thedynamic range of the A/D converter and digital baseband can be relaxedif the spectral estimation indicates that no interference exists.

In practice, interferers are detected through the wideband A/D bymeasuring a spectrum across and outside of the RX band. In particular,the signal out from the A/D is passed to a digital lowpass filter thatreduces the frequency range of detection (where crossmodulation andtypical worst case intermodulation is a concern) and to reducequantization noise of the A/D. This signal is then passed to a digitalhighpass filter that is the inverse of the analog low pass filter (320)to equalize the signal amplitude. This received signal spectrum can beprovided through a Fast Fourier Transform (FFT) or through a bank ofbandpass filters (BPF) covering the desired spectrum. The FFT size orBPF bandwidth is set by a desired frequency resolution, and the numberof points can be set by the desired frequency range. In addition, manyother techniques known in the art can be used to provide this spectrummeasurement. Additionally, other methods exist to equalize the signalamplitude such as simple addition of the inverse analog low pass filter(320) transfer function after spectral estimation. In this example, aset of twenty-four bandpass filters are used to cover the range of thespectrum to find the interference. The power levels of the interference(as well as the frequency offsets as will be detailed below) are used inby the DSP in equation 2, to calculate the proper linearity(predetermined empirically) of the receiver components to provide atleast a minimum SIR for proper reception by the communication device.Linearity is adjusted by either or both of a current control and gaincontrol to the mixer and/or amplifier, thereby mitigating the waste ofbattery current.

In a preferred embodiment, if the signal level in the various receivercomponents is reduced (i.e. lower signal swing), there is no need toprovide full dynamic range in the various components of the receiver toaccommodate this reduced signal. Therefore, in the present invention,the dynamic range of these components can be reduced to track thelinearity requirements of the LNA and mixer to further save current. Inparticular, current control can be used to control the dynamic range ofone or more of the analog filter block (320), A/D (322) and digitalfilter block (323).

FIG. 6 illustrates a method to mitigate the effects of third-orderintermodulation distortion, in accordance with the present invention.Referring to FIGS. 3-6, a first step 600 includes inputting the widebandsignal from the A/D 322. From this signal, a noise power spectrum isgenerated 602 using FFT or a bank of BPFs as described above. The noisepower spectrum can be determined under many conditions including duringperiods of non-communication with the receiver, during periods with thetransmitter on (or off), and during several periods using an average ofseveral measurements. A next step 604 includes detecting anyinterference products, which are found by measuring if any of the FFTbins or bandpass filters in the bank have a noise power level above apredetermined threshold indicating interference is present. Thepredetermined threshold is determined empirically from the finallinearity requirements of the receiver. The threshold can be a constantlevel or a curve tailored for the particular RX band. If none of thebins of filters present an interference noise power over the threshold,then interference is not considered a problem for device reception. Inthis case, the receiver (LNA 310 and mixer 318) can be set 606 to lowlinearity at their the lowest IP3 setting (and optionally set for lowdynamic range for the analog filter block 320, A/D 322 and digitalfilter block 323) using current and/or gain control.

However, if interference power is found over the threshold in any of thebins, a next step includes determining 608 a frequency offset of theinterferers. If the frequency offset (delta) between two interferers isa multiple, or particularly the same, as the delta between the nearestfrequency interferer and the RX band, then there will be third-orderinterference products in the RX band that will lower the SIR of thereceiver. If the frequency offsets are calculated 609 such thatinterference products will not fall in the RX passband, theninterference is not considered a problem for device reception, and thereceiver can be set 606 to low linearity (and optionally low dynamicrange) using current and/or gain control. However, if the frequencyoffsets are calculated 609 such that interference products fall in theRX passband, then it is necessary to calculate the receiver linearityrequired to achieve a desired signal-to-interference ratio and adjustthe receiver linearity calculated in the calculating step to achieve thedesired signal-to-interference ratio. This is done in equation 2 byfirst setting 607 P1 to the power level of the interferer closest to theRX band and P2 to the power level of the interferer farthest from the RXband. Next 605, using the known or measured values of the required SIR601 and the received desired signal P_(sig) 603, a third order interceptpoint (IP₃) threshold can be calculated from equation 2, to produce thedesired SIR, wherein the receiver (LNA 310 and mixer 318) can be set toan appropriate linearity (and optionally an appropriate dynamic rangefor the analog filter block 320, A/D 322 and digital filter block 323)using current and/or gain control. Increasing linearity can be achievedby either lowering a gain of the gain stage and mixer of the receiver,increasing current to the gain stage and mixer of the receiver, or both.This adjustment can also include the analog filter block.

The present invention is also applicable to the mitigation ofcrossmodulation interference products. FIG. 7 shows a graph illustratingthe effects of crossmodulation on a receiver (RX) band. In this example,a first interferer, A₂ cos(ω₂t), which is near to the RX bandcrossmodulates with leakage from the devices transmitter, A₁(t)cos(ω₁t), which results in wideband interference in the RX band. Thiscrossmodulation produces non-linear distortions within the RX band thatdegrades signal-to-interference performance (as well as third-orderproducts outside of the band). In particular, the interference fromdistortion isν_(cmd)(t)= 3/2A ₁ ²(t)A ₂ cos(ω₂t)  (3)and the signal-to-interference ratio is $\begin{matrix}{\frac{P_{sig}}{P_{cmd}} = \frac{P_{sig}}{{2P_{1}} + P_{2} - {2{IP}_{3}} + 6 + {AF} + {AM}}} & (4)\end{matrix}$where P_(sig) is the power of the desired in-band (RX) signal andP_(cmd) is the power of the interference. P₁ is the power level of thetransmitter, and P₂ is the power level of the interferer. In the presentinvention, the power and frequency offsets of the transmitter andinterferer are known or measured directly to determine their effectwithin the RX band. In addition, a percent amplitude modulation of thetransmitter can also be used to determine the proper linearity of thereceiver. Moreover, an attenuation factor is defined due to thebroadband nature of the crossmodulation as will be described below. Witha given signal-to-interference ratio (SIR) limit (as defined in thestandard), an IP₃ threshold can then be found to provide sufficient SIR.The components of the receiver can then have their linearity adjusted toprovide sufficient SIR.

Referring back to FIG. 5, crossmodulation mitigation can be providedusing spectral signal estimation combined with an attenuation factor(AF) to calculate the receiver linearity required, based on the measuredinterference and a desired SIR. The attenuation factor is predeterminedfor the receiver and defines measurement attenuation per offset from theRX band and relates to correction of out-of-band power measurements. Inparticular, it should be noted that the attenuation factor captures thefact that the crossmodulation decreases with the offset frequency of theinterference. This fact allows the reduction our current/linearity basedupon the knowledge of the frequency offset of the interference. If thereis no interference in the 2 to 7 MHz frequency range, then the receivercan operate with minimal linearity independent of the amount of TXsignal as long as the crossmodulation and/or the intermodulationrequirements are met.

The crossmodulation solution finds best application in a multimode GSMand WCDMA communication system or multimode AMPS and cdma2000communication systems. Multimode systems often involve the deployment ofthe two modes in the same frequency band thus leading to interferencethat produces crossmodulation.

As represented in FIG. 5, the DSP module (308 in FIG. 3) can also beused to detect crossmodulation distortion, in accordance with thepresent invention. In this case, the non-linear distortion that isproduced is wideband due to the crossmodulation of the wideband TXsignal. The DSP uses inputs of the known transmitter (TX) power,modulation and frequency along with the received signals level (P_(sig))as well as the required SIR. Then, given the measurements of theinterferer and knowledge of the transmitter, can calculate an IP₃threshold from equation 4, to produce the desired SIR. In particular,the present invention uses spectral estimation to decide if anysignificant crossmodulation products from wideband interference willfall inside the receive channel filter. The dynamic range of the A/Dconverter and digital baseband can be relaxed if the spectral estimationindicates that no interference exists

As before, interference is detected through the wideband A/D bymeasuring a spectrum across and outside of the RX band. In particular,the signal out from the A/D is passed to a digital lowpass filter thatreduces the frequency range of detection for crossmodulation and toreduce quantization noise of the A/D. This signal is then passed to adigital highpass filter that is the inverse of the analog low passfilter (320) to equalize the signal amplitude. This received signalspectrum can be provided through a Fast Fourier Transform (FFT) orthrough a bank of bandpass filters (BPF) covering the desired spectrum.The FFT size or BPF bandwidth is set by a desired frequency resolution,and the number of points can be set by the desired frequency range. Inaddition, many other techniques known in the art can be used to providethis spectrum measurement. Additionally, other methods exist to equalizethe signal amplitude such as simple addition of the inverse analog lowpass filter (320) transfer function after spectral estimation. In thisexample, a set of twenty-four bandpass filters are used to cover therange of the spectrum to find the interference. The power levels andfrequency offsets of the interferer and transmitter are used in by theDSP in equation 4, to calculate the proper linearity (predeterminedempirically) of the receiver components to provide at least a minimumSIR for proper reception by the communication device. Linearity isadjusted by either or both of a current control and gain control to themixer and/or amplifier, thereby mitigating the waste of battery current.

In a preferred embodiment, if the signal level in the various receivercomponents is reduced (i.e. lower signal swing), there is no need toprovide full dynamic range in the various components of the receiver toaccommodate this reduced signal. Therefore, in the present invention,the dynamic range of these components can be reduced to track thelinearity requirements of the LNA and mixer to further save current. Inparticular, current control can be used to control the dynamic range ofone or more of the analog filter block (320), A/D (322) and digitalfilter block (323).

FIG. 8 illustrates a method to mitigate the effects of crossmodulationdistortion, in accordance with the present invention. Referring to FIGS.3, 5, 7 and 8, a first step 800 includes inputting the wideband signalfrom the A/D 322. From this signal, a noise power spectrum is generated602 using FFT or a bank of BPFs as described above. The noise powerspectrum can be determined under many conditions including duringperiods of non-communication with the receiver, during periods with thetransmitter on (or off), and during several periods using an average ofseveral measurements. A next step 604 includes detecting anyinterference products, which are found by measuring if any of the FFTbins or bandpass filters in the bank have a noise power level above apredetermined threshold indicating interference is present. Thepredetermined threshold is determined empirically from the finallinearity requirements of the receiver. The threshold can be a constantlevel or a curve tailored for the particular RX band. The threshold canalso be different for intermodulation and crossmodulation problems. Ifnone of the bins of filters present an interference noise power over thethreshold, then interference is not considered a problem for devicereception. In this case, the receiver (LNA 310 and mixer 318) can be set606 for low linearity at their lowest IP3 setting (and optionally setfor low dynamic range for the analog filter block 320, A/D 322 anddigital filter block 323) using current and/or gain control.

However, if interference power is found over the threshold in any of thebins, a next step includes determining 808 a frequency offset of theinterferer, which is used in combination with the corresponding inputattenuation 801 from a predetermined lookup table 800 along with themeasured power. If the frequency offset of the interferer is closeenough to the RX band (e.g. within 7 MHz for WCDMA), then there can becrossmodulation interference products in the RX band that will lower theSIR of the receiver. If the interference products are calculated 809such that interference products will not fall in the RX passband, theninterference is not considered a problem for device reception, and thereceiver can be set 606 to low linearity (and optionally low dynamicrange) using current and/or gain control.

However, if the interference products are calculated 609 such thatinterference products will fall in the RX passband, then it is necessaryto calculate the receiver linearity required to achieve a desiredsignal-to-interference ratio and adjust the receiver linearitycalculated in the calculating step to achieve the desiredsignal-to-interference ratio. Due to the broadband nature of theinterference, the FFT bins and AF points are normalized 802 withreference to the maximum value of the highest power bank that exceedsthe noise spectrum threshold. Then the bin values are summed 803 todetermine a normalized total crossmodulation contribution, R, forequation 4, and P₂ and AF are set to the level of the maximuminterferer, i.e. normalized to maximum. Next 805, using the known ormeasured values of the required SIR 601, the received desired signalP_(sig) 603, and a TX power level and percent amplitude modulation 807,a third order intercept point (IP₃) threshold can be calculated fromequation 4, to produce the desired SIR, wherein the receiver (LNA 310and mixer 318) can be set to an appropriate linearity (and optionally anappropriate dynamic range for the analog filter block 320, A/D 322 anddigital filter block 323) using current and/or gain control. Increasinglinearity can be achieved by either lowering a gain of the gain stageand mixer of the receiver, increasing current to the gain stage andmixer of the receiver, or both.

It should be recognized that the techniques described above can berepeated periodically to update the operating condition of the devicebased on changes in the interference. In addition, the DSP module 308can adjust the gain stage, mixer, A/D and filter blocks differentlydepending on interference conditions and the capability of each element.

Advantageously, the present invention addresses non-linear distortionsdirectly and separately from noise and interference than is done in theprior art. This is accomplished with existing hardware eliminating theneed for additional circuitry hence saving space on the printed circuitboard and within the integrated circuits. The ever-increasingcapabilities of digital signal processor technology allows for thesimultaneous measurements and operation of different modes of thecommunication signal to provide seamless control with very small chipdie areas.

As can be seen from the foregoing, the present invention provides amethod and apparatus for reducing the current drain in a communicationdevice by making the elements in the receiver (amplifiers and downconverters) less linear or with less dynamic range when there is minimalinterference, and increasing linearity or dynamic range again when theinterference is significant. The present invention can alsoadvantageously be applied to control, data and paging communication inaddition to the traffic channel that is described above. During theseevents the receiver linearity can be adjusted in the same mannerdescribed previously.

The present invention involves taking advantage of the enhanced dynamicrange of a wideband A/D converter to provide a novel technique to detectinterference. Using techniques such as FFTs or a bank of band passfilters, the present invention allows the detection of the presence ofinterference, the power of interference, and the frequency offset of theinterference. The useful range of frequency offset detection is limitedby the sigma delta A/D noise floor and the attenuation of the analogbaseband filter. Simulation results demonstrate that signals in the 2 to7 MHz offset range, with amplitudes from −60 dBm to the compressionpoint, can be detected. This is very important since the 7 MHz point isthe frequency offset at which the crossmodulation is no longer an issuefor WCDMA due to the frequency offset from the receiver. Furthermore,narrowband intermodulation tests performed as part of the WCDMA and CDMAstandards both fall within this frequency offset range.

The use of a simple yet more sophisticated detection of the presentinvention adds two novel features. One, the receiver linearity of the RFelements can be adjusted based not just on a quality metric as in theprior art but on actual measurements of interference, both power andfrequency offset, and calculation of linearity requirements based on theinterference. Two, the use of this detector can be used to reduce thedynamic range of the A/D and digital filter based on the amount ofinterference present. This provides the most efficient use of thelinearity/dynamic range of the various receiver components.

Although the invention has been described and illustrated in the abovedescription and drawings, it is understood that this description is byway of example only and that numerous changes and modifications can memade by those skilled in the art without departing from the broad scopeof the invention. Although the present invention finds particular use inportable cellular radiotelephones, the invention could be applied to anymultimode wireless communication device, including pagers, electronicorganizers, and computers. Applicants' invention should be limited onlyby the following claims.

1. A method for reducing current drain in a communication device, themethod comprising the steps of: detecting an interferer outside of areceiver passband of the communication device; measuring power levelsand frequency offsets of the interferer and a transmitter of thecommunication device; determining whether crossmodulation productsexceed a noise spectrum threshold within the receiver passband,whereupon, if the crossmodulation products exceed the noise spectrumthreshold, calculating a receiver linearity required to achieve adesired signal-to-interference ratio; and adjusting the receiverlinearity calculated in the calculating step to achieve the desiredsignal-to-interference ratio.
 2. A method as recited in claim 1, whereinthe detecting and measuring steps includes estimating a signal spectrumof the interference products of the interferer and transmitter.
 3. Amethod as recited in claim 1, wherein the calculating step includes anormalization of the interference using an attenuation factor of thereceiver at the frequency offset.
 4. A method as recited in claim 1,wherein the adjusting step includes adjusting a dynamic range of thereceive in accordance with the adjusted receiver linearity.
 5. A methodas recited in claim 1, wherein the calculating step includes calculatinga third-order intercept point threshold to provide sufficientsignal-to-interference, and wherein the adjusting step includes settingat least one of the group of current and gain to the receiver at a levelsufficient to at least meet the third-order intercept point threshold.